Method and apparatus for frame alignment in discrete multitone transceivers

ABSTRACT

A method and apparatus for determining frame alignment in a discrete multitone transceiver by determining a rotation of the coefficients of the shortening channel impulse response for which inter symbol interference (ISI) is minimal. The ISI is determined by multiplying the average value of a transmitted discrete multitone symbol and multiplying it by the coefficients of the shortening channel impulse response in a given rotation.

FIELD OF INVENTION

[0001] The invention pertains to transceivers and modems. Moreparticularly, the invention pertains to asymmetric digital subscriberline (ADSL) modems used to achieve very high speed data communicationvia telephone networks.

BACKGROUND OF THE INVENTION

[0002] There is an ever present desire to maximize the speed of digitalcommunications via networks, and particularly telecommunicationnetworks, such as public telephone systems. Accordingly,telecommunication network providers now offer customers many options forcoupling to the telephone network in addition to the standard analogbased connection commonly referred to as POTS (Plain Old TelephoneSystem). Some of the options that are widely available are IntegratedServices Digital Network (ISDN) lines, T-1 lines, E-1 lines, digitalsubscriber lines (DSL) and asymmetric digital subscriber lines (ADSL).

[0003] ADSL's can provide very high data speeds, such as on the order ofseveral megabits per second, over a standard twisted wire pair. Unlikethe traditional data modems used for analog communication with atelephone central office via a twisted wire pair, ADSL requires modemsboth at the subscriber end and at the telephone company Central Officeend. Current ADSL systems employ discrete multitone (DMT) technology toimplement high bandwidth communications, such as for digital TVbroadcast, on demand video, high speed video-based internet access, workat home digital file transfer, teleconferencing, home shopping, andother information services over existing twisted wire pair telephonelines.

[0004] Several DMT standards have been promulgated. For instance, theInternational Telecommunications Union (ITU) has promulgated a standardfor ADSL that is commonly termed G.lite and which is set forth in ITU-Tspecification G.992.2, incorporated herein by reference. Anotherstandard, promulgated by ANSI, is commonly termed Heavy ADSL and is setforth in ANSI specification T1.413, issue 2, also incorporated herein byreference. In DMT communications, data is sent in frames. A frame iscomprised of a plurality of samples, each frame including data samplesand cyclic prefix samples. Data samples comprise most of the frame andthe collection of data samples in a single frame comprise one DMTsymbol. The cyclic prefix is added at the beginning of each frame andcomprises the last L samples of that frame. Accordingly, the cyclicprefix samples are between the DMT symbols in the data stream. Thepurpose of the cyclic prefix is to help avoid inter-symbol interference(ISI). The frame and cyclic prefix is of a standardized length. Forexample, in heavy ADSL, each symbol comprises 512 samples with 256 tones(32 tones for upstream communications), each tone having a real and animaginary portion. Heavy ADSL utilizes a cyclic prefix of length L=32samples. Accordingly, a frame has 544 samples.

[0005]FIG. 1 is a block diagram of the basic ADSL modem functions, aswould be well known to those of skill in the art. The upper half of thediagram represents functions in the transmit direction while the lowerhalf represents functions in the receive direction.

[0006] It should be noted that FIG. 1 is a functional block diagram andthat the blocks shown therein do not necessarily correspond to separatephysical circuits. In fact, most if not all of the functions will beperformed by one or more digital processors such as, but not limited to,a digital signal processor, a micro processor, a programmed generalpurpose computer, etc. It also is possible that part or all of thefunctions of some of the blocks may be implemented by analog circuitry.

[0007] In the transmit direction, digital data is transmitted from thetransmitter 102 to a scrambler 104 that scrambles the data fortransmission. The data is then processed through a forward errorcorrection (FEC) encoder 106 which adds syndrome bytes to the data. Thesyndrome bytes will be used for error correction by the receiver at thereceiving terminal. Next, as shown in block 108, the transmit data isencoded using quadrature amplitude modulation (QAM). The data is thenconverted from the frequency domain to the time domain via Inverse FastFourier Transform (IFFT) 110.

[0008] A 1:4 interpolator 112 interpolates the output of IFFT block 110to produce 512 samples from the 128 samples output from block 110. The32 sample cyclic prefix is added to each frame in block 114. The data isthen forwarded to a coder/decoder (CODEC) 116. The CODEC encodes thedata for transmission over the twisted wire pair to the receivingdevice.

[0009] In the receiver portion of transceiver 100, the received signalis passed from the twisted wire pair through the CODEC 116 where it isdecoded. It is then passed to a time domain equalizer (TEQ) 118 toshorten the channel impulse response. Then, in 120, the cyclic prefix isremoved. It will be appreciated by those of skill in the related artsthat, if the length of the channel impulse response is less than orequal to the cyclic prefix length, then the Inter-Symbol Interference(ISI) can be eliminated by removing the Cyclic Prefix (CP) length.Furthermore it is possible to compensate for channel distortion with thefrequency domain equalization (FEQ) in block 124, discussed furtherbelow.

[0010] An echo canceller (EC) 134 is coupled between the transmit pathand the receive path and creates an echo cancellation signal based onthe transmit signal which is subtracted by subtractor 121 from thereceive signal in order to cancel any echo of the transmit signal thatis present on the twisted wire pair that could interfere with thereceived signal. The residual echo signal is converted back to thefrequency domain by fast fourier transform (FFT) in block 122.

[0011] The received signal, which now has had the cyclic prefix removedand has been converted back to the frequency domain is sent to block124, where frequency domain equalization (FEQ) is employed to compensatefor the channel distortion.

[0012] The received signal is then processed through a quadratureamplitude modulation (QAM) decoder 126 to decode the tone signal intodigital data. That is followed by forward error correction (FEC) inblock 128 which uses the syndrome bits that were added by the transmitpath FEC encoder 106 to perform forward error correction. Finally, thedata is descrambled in block 130 to extract the true data signal andthen forwarded to a receiver 132.

[0013] The output of the FFT block 122 also is sent to a timing recoverycircuit 136 that controls the CODEC 116 to synchronize the CODEC to thetiming of the received data. Essentially, the timing recovery process isa feedback process in which the timing tones are detected and used tocontinuously adjust the CODEC timing so as to sample the received dataat the appropriate sampling points.

[0014] In prior art frame alignment schemes, during initialization, thetransmitter transmits a known pattern in a frame and the receiverattempts to receive that pattern. More particularly, the transmittertransmits the same pattern in a Who plurality of frames, for example1024 consecutive frames. The receiver performs a cross-correlation ofthe data on the two wire pair with the expected pattern while thetransmitter transmits those 1024 frames and determines which set of L+1consecutive samples yields the peak cross correlation. The timing thatresulted in the peak cross correlation calculation is selected as thestart time of a frame or a symbol. It means that the sample thatresulted in the peak cross correlation calculation is the first sampleof the frame, e.g., 544 samples (before removing the cyclic prefix).

[0015] This technique for frame alignment or symbol alignment iscomputationally demanding and is not as accurate as desired.

[0016] Accordingly, it is an object of the present invention to providean improved method and apparatus for frame alignment in a DMTtransceiver.

[0017] It is another object of the present invention to provide animproved DMT transceiver.

SUMMARY OF THE INVENTION

[0018] The invention is a method and apparatus for aligning with atransmit frame in a discrete multitone (DMT) transceiver so as tominimize inter-symbol interference (ISI). The method and apparatusinvolves calculating the shortening channel impulse response and thendetermining the particular set of consecutive samples of length L+1 inthe shortening channel impulse response, that has a maximal energy. Oncethis particular set of coefficient is determined, the first coefficientof that set is selected as a starting location h′₀ in shortening impulseresponse for determining frame alignment. This can be achieved byskipping samples of the receive signal in a particular frame.

[0019] The majority of the interference of the immediately precedingsymbol on the present symbol typically is approximately contributed byshortening channel impulse response coefficients h′_(L+1), h′_(L+2), . .. h′_(P−1), where P is the DMT symbol length (e.g., 512 samples). Thisis true because the channel impulse response coefficients, h′₀, h′₁, . .. h′_(L), do not contribute to the ISI. Therefore, the majority of theinterference can be calculated with the shortening channel impulseresponse coefficients h′_(L+1), h′_(L+2), - - - and h′_(P−1).

[0020] The noise on the i-th subcarrier is a complex value r_(i)+s_(i)j,where j={square root}{square root over (−1)}, and the absolute value{square root}{square root over (r_(i) ²+s_(i) ²)} is considered to bethe ISI noise on the i-th subcarrier. The total ISI noise is computed bysumming all the inner products of the weight factor vector W and the ISIleakage energy vector from the previous symbol, {overscore(C)}(|h_(i)′|FV_(k). The rotation of the shortening channel impulseresponse, that yields the smallest total ISI noise will be chosen as theframe alignment point.

[0021] Accordingly, to determine the best rotation of the shorteningchannel impulse response coefficients for frame alignment, we calculatethe ISI using the rotation as described above as well as severalimmediately surrounding rotations of the coefficients and select therotation that yields the lowest ISI value. The calculations areperformed in the frequency domain. Specifically, we multiply the averageenergy of the transmitted symbol (which is a constant) by the FastFourier Transform (FFT) of the ISI value caused by the imperfectshortening channel impulse response coefficients. We do this for variousrotations including and immediately surrounding the rotation determinedabove and take the rotation that yields the smallest value. It has beenfound that excellent results can be achieved by calculating theinterference value for a relatively small number of rotations. Forinstance, in a DMT system with 512 samples per symbol and a 32 samplecyclic prefix, excellent results can be achieved by calculatinginterference for about seven different rotations surrounding andincluding the rotation corresponding to the maximum channel response.

BRIEF DESCRIPTION OF THE DRAWINGS

[0022]FIG. 1 is a block diagram illustrating the functions of an ADSLmodem transceiver in accordance with the prior art.

[0023]FIG. 2 is a block diagram illustrating in more detail channelshortening in discrete multitone communications in accordance with theprior art.

[0024]FIG. 3 is a block diagram illustrating the functions of an ADSLmodem transceiver for digital multitone communications in accordancewith the present invention.

[0025]FIG. 4 is a flowchart illustrating symbol alignment in accordancewith the present invention.

DETAILED DESCRIPTION OF THE INVENTION

[0026]FIG. 2 is a block diagram that helps illustrate time domainequalization in DMT in more detail. In particular, block 203 encompassesfunctions performed at the transmitter. Block 207 encompasses functionsthat occur in the data channel and block 213 encompasses functionsperformed at the receiver. Block 205 encompasses those functions whichresult in channel shortening. Thus, it can be seen that, in thetransmitter 203, a DMT signal, x(n), is generated for transmission viathe channel. The channel 207 has a channel impulse response, h(n),represented by block 209. Accordingly, the signal, y(n), received at thereceiver 213 is the convolution of the original transmitted signal x(n)and the channel response h(n), i.e.;

y(n)=x(n)*h(n)  Eq. (1)

[0027] The time domain equalization circuit (TEQ) 211 in the receiver213 is a short time-domain finite impulse response (FIR) filter thatconvolves the received signal, y(n), with a coefficient 1+a(n). Theoutput y′(n) from the time domain equalization domain circuit 211therefore is:

y′(n)=y(n)*(1+a(n))=x(n)*(h(n)*(1+a(n)))=x(n)*h′(n)  Eq. (2)

[0028] Let us call h(n)*(1+a(n)) the shortening channel impulse response(or shortening channel) and denote it as h′(n) having a length M.

[0029] The TEQ circuit 211 (i.e., Eq. (2)) shortens the channel impulseresponse to the length of the cyclic prefix, e.g., 32 samples. Then thecyclic prefix is removed from the signal y′(n) in block 215.

[0030] If the shortening channel length, M, is less than or equal to thelength, L, of the cyclic prefix, i.e., M≦L, then each y′(n) will have nodependence on any transmitted samples before x(n−L), i.e., there will beno inter-symbol interference (ISI).

[0031] Thus, after the cyclic prefix is removed, the actual symbol willremain without inter symbol interference. However, if M>L, then thesampling signal y′(n) is dependent on the samples before x(n−L) and,thus, the removal of the cyclic prefix would not prevent inter-symbolinterference and thus would degrade the performance of the transceiver.Hence, it is desirable to design an efficient TEQ shortening filter,1+a(n), that makes M≦L and for which the energy in the channel impulseresponse h′(n) with 0≦n<512 is as small as possible for n≧L and as largeas possible for n<L.

[0032] For proper operation of the transceiver in accordance with thedescription above, of course, the frame in the receiver must alignitself to the frame in the remote transmitter in order to ‘i’_read theframe samples correctly. The proper frame alignment point is determinedduring initialization.

[0033] U.S. patent application Ser. No. 09/481,027 entitled “Method andApparatus for Time-Domain Equalization in Discrete MultitoneTransceivers”, filed on Jan. 11, 2000, describes a TEQ shortening filterthat produces an overall channel response h′(n) for which M≦L, asdesired in order to eliminate ISI.

[0034] The impulse response h(n) of the channel has the transferfunction: $\begin{matrix}{{H(z)} = {\frac{B(z)}{A(z)} = \frac{{b_{0} + {b_{1}z^{- 1}} + {b_{2}z^{- 2}} + \ldots + {b_{({L - 1})}z^{- {({L - 1})}}}}\quad}{1 + {a_{1}z^{- 1}} + {a_{2}z^{- 2}} + \ldots + {a_{({K - 1})}z^{- {({K - 1})}}}}}} & \text{Eq. (3)}\end{matrix}$

[0035] where L is the length of the cyclic prefix, and K is the lengthof the TEQ filter having impulse response (1+a(n)) or A(z). Inaccordance with equation (3), the transfer function of the shorteningchannel is:

{overscore (h)}(z)=A(z)H(z)=b ₀ +b ₁ z ⁻¹ +b ₂ z ⁻² + . . . +b _((L−1))z ^(−(L−1))  Eq. (4)

[0036] Thus, the length, M, of the shortening channel is set equal tothe length of the cyclic prefix, L, so that ISI between two consecutivesymbols can be eliminated to improve the overall performance of thetransceiver by removing the cyclic prefix.

[0037] Let

I=[b₀ b₁ b₂ . . . b_((L−I)) −a, −a₂ . . . a_((K−1))]^(T)=[b a]^(T)  Eq.(5)

[0038] and

W(n)=[x(n) x(n−1) x(n−2) . . . x(n−(L−1))y(n−1) y(n−2) . . .y(n−(K−1))]^(T)  Eq. (6)

[0039] where T denotes the transpose of a matrix. From equations (3)through (6), the estimated signal, {tilde over (y)}(n), at the receiveris:

{tilde over (y)}(n)=I ^(T) W(n)  Eq. (7)

[0040] and the error is:

e(n)=y(n)−{tilde over (y)}(n)  Eq. (8)

[0041] From the well known LMS (Least Mean Square) error algorithm, theoptimal solution I_(opt) is

RI _(opt) =r  Eq. (9)

[0042] where

R=E{W(n)W ^(T)(n)}  Eq. (10)

r=E{y(n)W(n)}  Eq. (11)

[0043] E{ } in the equations above represents a generalized averagingfunction or an ensemble average.

[0044] Since some channel interference is caused by channel noise andimpairments, it is very difficult to achieve a shortening channelimpulse response that has only (L+1) continuous nonzero values and zerosin the remaining slots. Hence, accurate frame alignment can be verycritical to the overall performance of DMT transceivers.

[0045] Let [h₀′, h₁′, h₂′, . . . h₅₁₁′] be the shortening channelimpulse response. As previously mentioned, one way to locate the framealignment roughly is by finding the series of consecutive coefficientsof length L+1 (L is the channel prefix length) that yields the maximalenergy. By skipping samples in the receive signal in a particular frame,we can allocate the first coefficient of the series of consecutivecoefficients of length L+1 as h′₀, whereby the coefficients from h′₀ toh′_(L) yield the maximal energy. The coefficients of the shorteningchannel impulse response from h′_(L+1) to h′_(P−1) contribute to lSl.Hence, we try to make the energy as low as possible for h′_(L+1) toh′_(P−1). Although there is maximal energy in h′₀ to h′_(L), it is notnecessarily the best choice for frame alignment. Rather, we know onlythat the frame alignment point is close to this point. Thus, weimplement a fine-tuning algorithm in accordance with the presentinvention to find the best location for frame alignment.

[0046] Let c=[c₀, c₁, c₂, . . . c₅₁₁] and d=[d₀, d₁,d₂, . . . d₅₁₁] bethe samples of previous symbol and current symbol, respectively, thatare sent from the transmitter. Due to the addition of the 32 samples ofcyclic prefix, the samples that go out through the CODEC and channelwill be [c₀, c₁, c₂, . . . c₅₁₁, d₄₈₀, d₄₈₁, . . . d₅₁₁, d₀, d₁, d₂, . .. d₅₁₁]. Let the corresponding samples at the receiver be [c′₀, c′₁,c′₂, . . . c′₅₁₁, d″₄₈₀, d″₄₈₁, . . . d″₅₁₁, d′₀, d′₁, d′₂, . . .d′₅₁₁]. Thus, after removing the cyclic prefix, we have:

d′ ₀ =d ₀ h′ ₀ +d ₅₁₁ h′ ₁ +d ₅₁₀ h′ ₂ + . . . +d ₄₈₀ h′ ₃₂ +c ₅₁₁ h′₃₃ + . . . +c ₃₃ h′ ₅₁₁  (12)

d′ ₁ =d ₁ h′ ₀ +d ₀ h′ ₁ +d ₅₁₁ h′ ₂ + . . . +d ₄₈₀ h′ ₃₃ +c ₅₁₁ h′ ₃₄ +. . . +c ₃₂ h′ ₅₁₁  (13)

d′ ₅₁₁ =d ₅₁₁ h′ ₀ +d ₅₁₀ h′ ₁ +d ₅₀₉ h′ ₂ + . . . +d ₀ h′ ₅₁₁  (14)

[0047] From equations (12) to (14) and the imperfectly shorteningchannel impulse response, the previous symbol will interfere with thecurrent symbol. The interference is c₅₁₁h′₃₃+ . . . +c₃₃h′₅₁₁ in d′₀,c₅₁₁h′₃₄+ . . . +c₃₂h′₅₁₁ in d′₁, etc. If we assume that the transmittedsignal is a pseudo random and stationary signal, the major interferencefrom the previous symbol on the current symbol is h′₃₃, . . . , h′₅₁₁.That is: $\begin{matrix}{{{{{INTF} = {{\overset{\_}{C}\left\lbrack \left| h_{33}^{\prime} \middle| {+ \left| h_{34}^{\prime} \middle| {+ \left| h_{35}^{\prime} \middle| {{+ \ldots} +} \middle| h_{511}^{\prime} \middle| {\quad,} \right.} \right.} \right.\quad  \right.}h_{34}^{\prime}{ + }h_{35}^{\prime}}}}\quad} + {{h_{36}^{\prime}{{{+ \ldots} +}}h_{511}^{\prime}{{,\ldots \quad,}\quad }h_{511}^{\prime}\left. \underset{\underset{33}{}}{{,0},\ldots \quad,0} \right\rbrack}\quad}} & (15)\end{matrix}$

[0048] where {overscore (C)} is the average energy value of thetransmitted symbol and INTF represents the symbol interference. Equation(15) can be rewritten as: $\begin{matrix}{{INTF} = {\overset{\_}{C}\left( \left| h_{33}^{\prime} \middle| {\left\lbrack \overset{\overset{512}{}}{{1,\quad 0,\quad 0},\ldots \quad,0} \right\rbrack + h_{34}^{\prime}} \middle| {\overset{\overset{512}{}}{\left\lbrack {1,\quad 1,\quad 0{,\quad.\quad.\quad.\quad,}\quad 0} \right\rbrack} +} \middle| h_{35}^{\prime} \middle| {\left\lbrack {{1,\quad 1,\quad 1,\quad 0},{\ldots \quad,\quad 0}} \right\rbrack + \ldots} \middle| h_{511}^{\prime} \middle| \overset{\overset{512}{}}{\left\lbrack {\underset{\underset{479}{}}{1,\quad 1,\quad 1{,\quad.\quad.\quad.\quad {,\quad}}}\left. \left. {1,\underset{\underset{33}{}}{0,\ldots \quad,\quad 0}} \right\rbrack \right)} \right.} \right. \right.}} & (16)\end{matrix}$

[0049] In equation (16), {overscore (C)} is a constant and can beignored. The coefficient of the h′₅₁₁ shortening channel impulseresponse contributes more symbol interference that the coefficient h′₃₃and the coefficients from h′₀ to h′₃₂ do not affect the symbolinterference at all due to the removal of the cyclic prefix.

[0050] In accordance with the analysis above, the symbol interference ofthe previous symbol on the current symbol is caused by the nonzerocoefficients in the shortening channel impulse response. In order toreduce the symbol interference, we rotate the shortening channel impulseresponse [h′₀, h′₁, h′₂, . . . , h′₅₁₁] such that the last 479 (512−33)coefficients of the shortening channel impulse response have less symbolinterference on the next symbol.

[0051] Let us define V_(k) as$V_{k} = {\left\lbrack {\underset{\underset{kls}{}}{1,1,\ldots \quad,1},\quad \underset{\underset{{({512 - k})}0s}{}}{0,\ldots \quad,}} \right\rbrack \quad.}$

[0052] Then equation (15) and (16) can be rewritten as $\begin{matrix}\begin{matrix}{{INTF} = \left\lbrack {{{intf}_{0},{intf}_{1}},\ldots \quad,\quad {intf}_{511}} \right\rbrack} \\{= {\overset{\_}{C}\left( \left| h_{33}^{\prime} \middle| {V_{1} +} \middle| h_{34}^{\prime} \middle| {V_{2} + \ldots +} \middle| h_{511}^{\prime} \middle| V_{479} \right. \right)}}\end{matrix} & (17)\end{matrix}$

[0053] We can determine how much effect the symbol interference has oneach tone in the receiver by taking the FFT (Fast Fourier Transform) ofequation (16), as follows; $\begin{matrix}\begin{matrix}{{FINTF} = {{FFT}_{512}\left( \left\lbrack {{intf}_{0},\quad {intf}_{1}{,\quad.\quad.\quad.\quad,}\quad {intf}_{511}} \right\rbrack \right)}} \\{= {\overset{\_}{C}\left( \left| h_{33}^{\prime} \middle| {{FV}_{1} +} \middle| h_{34}^{\prime} \middle| {{FV}_{2} + \quad {.\quad.\quad.\quad +}} \middle| h_{511}^{\prime} \middle| {FV}_{479} \right. \right)}}\end{matrix} & (18)\end{matrix}$

[0054] where

[0055] FFT₅₁₂(V_(k))=(r₀+s₀j, r₁+s₁j, . . . , r₂₅₅+s₂₅₅j) and

[0056] FV_(k)=[t₀, t₁, . . . , t₂₅₅] is a 256 real value vector witht_(i)={square root}{square root over (r_(i) ²+s_(i) ²)}.

[0057] In ADSL, the different tones will be loaded by different numbersof bits, depending on the tone's SNR (Signal to Noise Ratio). The noiseis caused by ISI and all other effects of line impairments (such asecho, noise from radio signals, etc.) A bit loading algorithm willassign a certain number of bits to each tone based on the SNR, thecapacity of bit rate, and other factors. For more generality, inequation (18), the noise on each tone can be weighted by a differentvalue, W₀, W₁, . . . , W₂₅₅. In that case, the weight factor vector W onthe 256 tones would be the vector.

W=[W₀, W₁, . . . , W₂₅₅]^(T)  (19)

[0058] Then the total noise, TN, caused by ISI can be expressed as$\begin{matrix}\begin{matrix}{{TN} = {\left| {{total}\quad {ISI}\quad {noise}} \right| = \left| {{FINTF} \cdot W} \right|}} \\{= {\overset{\_}{C}\left( \left| h_{33}^{\prime} \middle| {\left( {{FV}_{1} \cdot W} \right) +} \middle| h_{34}^{\prime} \middle| {\left( {{FV}_{2} \cdot W} \right) + \quad {.\quad.\quad.\quad +}} \middle| h_{511}^{\prime} \middle| \left( {{FV}_{479} \cdot W} \right) \right. \right)}}\end{matrix} & (20)\end{matrix}$

[0059] where

[0060] |x| denotes the absolute value of x,

[0061] FV_(i) is a vector of 256 real values, and

[0062] FV_(i)·W is an inner product of FV_(i) and W.

[0063] The weight factors W_(i) in equation (19) could be different foreach of the 256 tones, depending on the importance of the tone based onits location. For example, the weight factors for the tones from 0 to 31can be set to zeros since they are used for transmitted signals. Also,the weight factor for the timing tone (e.g., W₆₃) can be set slightlyhigher than the weight factor of the other tones since tone 63 is animportant tone because it is involved in the performance of timingrecovery. If timing recovery performance is poor, it will degrade theoverall performance of the system and affect all of the tones.

[0064] Equation (20) exactly describes the symbol interference by thenonzero coefficients h′₃₃, . . . , h′₅₁₁ in the shortening channelimpulse response on each tone in the receiver. Since the first 33samples, from h₀′ to h₃₂′, in the shortening channel impulse response donot cause symbol interference, we can start to find the maximal energyof L+1 (33) consecutive coefficients of the shortening channel impulseresponse by rotating the shortening channel impulse responsecoefficients in a particular frame such that the coefficients from h₀′to h₃₂′, have the maximal energy. After that, we employ equation (20) tofine-tune to the best rotation around this range, i.e., the rotationthat yields the minimal TN value.

[0065] For instance, if we want to consider this rotation plus the threerotations to each side of this rotation (for a total of seven testedrotations), then we would perform equation (20) a total of seven times,as follows:${TN}_{1} = {\left| {{FINTF}_{1} \cdot W} \right| = {\overset{\_}{C}\left( \left| h_{30}^{\prime} \middle| {\left( {{FV}_{1} \cdot W} \right) +} \middle| h_{31}^{\prime} \middle| {\left( {{FV}_{2} \cdot W} \right) +} \middle| h_{32}^{\prime} \middle| {\left( {{FV}_{3} \cdot W} \right) + \quad {.\quad.\quad.\quad +}} \middle| h_{507}^{\prime} \middle| {\left( {{FV}_{478} \cdot W} \right) +} \middle| h_{508}^{\prime} \middle| \left( {{FV}_{479} \cdot W} \right) \right. \right)}}$${TN}_{2} = {\left| {{FINTF}_{2} \cdot W} \right| = {\overset{\_}{C}\left( \left| h_{31}^{\prime} \middle| {\left( {{FV}_{1} \cdot W} \right) +} \middle| h_{32}^{\prime} \middle| {\left( {{FV}_{2} \cdot W} \right) +} \middle| h_{33}^{\prime} \middle| {\left( {{FV}_{3} \cdot W} \right) + \quad {.\quad.\quad.\quad +}} \middle| h_{508}^{\prime} \middle| {\left( {{FV}_{478} \cdot W} \right) +} \middle| h_{509}^{\prime} \middle| \left( {{FV}_{479} \cdot W} \right) \right. \right)}}$${TN}_{3} = {\left| {{FINTF}_{3} \cdot W} \right| = {\overset{\_}{C}\left( \left| h_{32}^{\prime} \middle| {\left( {{FV}_{1} \cdot W} \right) +} \middle| h_{33}^{\prime} \middle| {\left( {{FV}_{2} \cdot W} \right) +} \middle| h_{34}^{\prime} \middle| {\left( {{FV}_{3} \cdot W} \right) + \quad {.\quad.\quad.\quad +}} \middle| h_{509}^{\prime} \middle| {\left( {{FV}_{478} \cdot W} \right) +} \middle| h_{510}^{\prime} \middle| \left( {{FV}_{479} \cdot W} \right) \right. \right)}}$${TN}_{4} = {\left| {{FINTF}_{4} \cdot W} \right| = {\overset{\_}{C}\left( \left| h_{33}^{\prime} \middle| {\left( {{FV}_{1} \cdot W} \right) +} \middle| h_{34}^{\prime} \middle| {\left( {{FV}_{2} \cdot W} \right) +} \middle| h_{35}^{\prime} \middle| {\left( {{FV}_{3} \cdot W} \right) + \quad {.\quad.\quad.\quad +}} \middle| h_{510}^{\prime} \middle| {\left( {{FV}_{478} \cdot W} \right) +} \middle| h_{511}^{\prime} \middle| \left( {{FV}_{479} \cdot W} \right) \right. \right)}}$${TN}_{5} = {\left| {{FINTF}_{5} \cdot W} \right| = {\overset{\_}{C}\left( \left| h_{34}^{\prime} \middle| {\left( {{FV}_{1} \cdot W} \right) +} \middle| h_{35}^{\prime} \middle| {\left( {{FV}_{2} \cdot W} \right) +} \middle| h_{36}^{\prime} \middle| {\left( {{FV}_{3} \cdot W} \right) + \quad {.\quad.\quad.\quad +}} \middle| h_{511}^{\prime} \middle| {\left( {{FV}_{478} \cdot W} \right) +} \middle| h_{0}^{\prime} \middle| \left( {{FV}_{479} \cdot W} \right) \right. \right)}}$${TN}_{6} = {\left| {{FINTF}_{6} \cdot W} \right| = {\overset{\_}{C}\left( \left| h_{35}^{\prime} \middle| {\left( {{FV}_{1} \cdot W} \right) +} \middle| h_{36}^{\prime} \middle| {\left( {{FV}_{2} \cdot W} \right) +} \middle| h_{37}^{\prime} \middle| {\left( {{FV}_{3} \cdot W} \right) + \quad {.\quad.\quad.\quad +}} \middle| h_{0}^{\prime} \middle| {\left( {{FV}_{478} \cdot W} \right) +} \middle| h_{1}^{\prime} \middle| \left( {{FV}_{479} \cdot W} \right) \right. \right)}}$${{TN}_{7} = {\left| {{FINTF}_{7} \cdot W} \right| = {\overset{\_}{C}\left( \left| h_{36}^{\prime} \middle| {\left( {{FV}_{1} \cdot W} \right) +} \middle| h_{37}^{\prime} \middle| {\left( {{FV}_{2} \cdot W} \right) +} \middle| h_{38}^{\prime} \middle| {\left( {{FV}_{3} \cdot W} \right) + \quad {.\quad.\quad.\quad +}} \middle| h_{0}^{\prime} \middle| {\left( {{FV}_{478} \cdot W} \right) +} \middle| h_{1}^{\prime} \middle| \left( {{FV}_{479} \cdot W} \right) \right. \right)}}}~$

[0066] The one of these seven values for TN that has the smallest valueis selected as the best rotation to use for frame alignment. Forexample, if TN₄ is the minimal ISI, then there is no need to do anythingfurther because the original frame alignment is the best one. However,if TN₃ is the minimal ISI, then the receiver should skip 511 (i.e.,512−1) samples in one frame in order to rotate the shortening channelimpulse response as follows:

[0067] h′₀→h′₁,h′₁→h′₂, . . . , h′₅₁₁→h′₀.

[0068] In practical implementations, the FV_(i)·W computations can bestored in tables to reduce processing time.

[0069]FIG. 3 is a block diagram of a DMT transceiver in accordance withthe present invention. Since the present invention pertains to thedetermination of proper frame alignment during initialization, all ofthe blocks shown in prior art FIG. 1 remain the same and are shown inFIG. 3 bearing the same reference numerals. The invention residesprimarily in frame alignment blocks 140 and 146 and switches 142 and144. Particularly, the processing described above is performed duringinitialization on the received data. Switches 142 and 144 are set tocause the received data to bypass blocks 140 and 146 during normaloperations.

[0070] In operation, block 140 first cross-correlates the received datawith a pre-defined periodic transmitted pattern frame signal to find thereal channel impulse response without shortening the channel. Thisprocess results in a coarse location of the frame alignment position andprepares for TEQ training. TEQ training is then performed. After thecompletion of TEQ training, the shortening channel impulse response isdetermined in block 146 in accordance with equation 20 to fine-tune thecorrect sampling position for final frame alignment.

[0071] The process performed in block 146 is illustrated in the flowchart of FIG. 4. The process starts at step 400. In step 402, theshortening channel impulse response h′, is determined in accordance withwell known mathematical steps. Then, in step 404, the shortened impulseresponse is queried to find the set of L+1 consecutive samples that hasthe maximum energy. Variable i is then set to the number of the firstsample of that set of samples (i.e., the shortened channel impulseresponse is rotated to the station for which the first).

[0072] In step 406, counter n is set to 1 and i is set to i−3. Note thatthe value of i must be calculated modulo P, where P is the shorteningchannel impulse response length, e.g., 512 for heavy ADSL. For example,if, in step 404, i is set to 2, then i−3 is 511 when counting modulo512. In step 408, equation (20) is used to calculate TN for thisparticular rotation of the samples. The result is saved as TN_(n). Inthe example that is illustrated in FIG. 4, TN will be calculated seventimes, namely, for those locations starting at i−3, i−2, i−1, i, i+1,i+2, and i+3. Accordingly, in step 412, it is determined whether TN hasbeen calculated for all seven rotations. If not, flow proceeds to step410 where both i and n are incremented and flow returns to step 408where TN is calculated for the next rotation.

[0073] When it is determined in step 412 that TN has been calculated forall seven rotations, flow proceeds to step 414. In step 414, therotation that yielded a minimum value for TN (for the seven rotationstested) is selected as the rotation to be used for frame alignmentduring normal operation. The process ends at step 416.

[0074] It will be understood by those of skill in the art that n can beset to any other reasonable value. It has been found that the bestrotation for frame alignment (as determined by equation 20) typicallywill be within about three samples of the rotation corresponding to themaximal channel impulse response energy in L+1 consecutive coefficientsas determined in step 404. Accordingly, n=7 is a reasonable choice.However, it can be any number less than 512. For general application,the weighed vector W in equation 19 can be set to$\underset{\underset{32}{}}{\left\lbrack {0,\ldots \quad,0} \right.}{\quad,}\quad {\underset{\underset{480}{}}{\left. {1,\ldots \quad,1} \right\rbrack}\quad.}$

[0075] However, it can be set to provide a different value for each toneif desired.

[0076] Having thus described a few particular embodiments of theinvention, various alterations, modifications, and improvements willreadily occur to those skilled in the art. Such alterations,modifications and improvements as are made obvious by this disclosureare intended to be part of this description though not expressly statedherein, and are intended to be within the spirit and scope of theinvention. Accordingly, the foregoing description is by way of exampleonly, and not limiting. The invention is limited only as defined in thefollowing claims and equivalents thereto.

I claim:
 1. A method of generating a shortening channel impulse response in a discrete multitone transceiver, said method comprising the steps of: (1) determining an impulse response of a channel, said impulse response having a plurality of coefficients corresponding to a length of a symbol; (2) rotating said impulse response coefficients to a rotation that decreases inter-symbol interference.
 2. The method of claim 1 wherein step (2) comprises rotating said impulse response coefficients to a rotation in which the first L+1 coefficients of said channel impulse response is maximal, where L is a length of said cyclic prefix.
 3. The method of claim 1 wherein step (2) comprises rotating said impulse response coefficients to a rotation that starts with coefficient L+1, where L is a length of said cyclic prefix.
 4. The method of claim 1 wherein step (2) comprises the steps of: (2.1) selecting a plurality of rotations of said channel impulse response including and surrounding said rotation that starts with coefficient L+1, where L is a length of said cyclic prefix; (2.2) calculating a value for inter-symbol interference based on each of said rotations; and (2.3) selecting a one of said rotations of said channel impulse response that yields the lowest inter-symbol interference value.
 5. The method of claim 4 wherein step (2.2) is performed in the frequency domain.
 6. The method of claim 5 wherein step (2.2) comprises: (2.2.1) generating Fourier transforms of said coefficients of said channel impulse response; (2.2.2) calculating an average value of a transmitted discrete multitone symbol; and (2.2.3) multiplying said Fourier transforms of said coefficients with said average.
 7. The method of claim 6 wherein, in step (2.2.1), said Fourier transforms are generated by fast Fourier transform.
 8. The method of claim 3 wherein step (2.2) comprises calculating FINTF={overscore (C)}(|h′ _(Y)|(FV ₁ ·W)+|h′ _(Y+1)|(FV ₂ ·W)+ . . . +|h′ _(P−1)|(FV _(P−Y) ·W)) where {overscore (C)}=an average value of a transmitted discrete multitone symbol, Y=an integer selected based on the number of the next to last coefficient of the set of consecutive coefficients determined in step (3.1), P=the number of coefficients in said shortening channel impulse response, and W=is a weighting factor vector [w₀, w₁, w₂ . . . , w_(P−1)]^(T).
 9. The method of claim 8 wherein w₀, w₁, . . . , w_(L)=0 and w_(L+1), w_(L+2), . . . , w_(P−1)=1.
 10. A method of frame alignment in a discrete multitone transceiver, said method comprising the steps of: (1) determining an impulse response of a channel, said impulse response having a plurality of coefficients corresponding to a length of a symbol; (2) rotating said impulse response coefficients to a rotation that decreases inter-symbol interference value; and (3) using said rotation for frame alignment.
 11. The method of claim 10 wherein step (2) comprises rotating said impulse response coefficients to a rotation in which the first L+1 coefficients of said channel impulse response is maximal, where L is a length of said cyclic prefix.
 12. The method of claim 10 wherein step (2) comprises the steps of: (2.1) selecting a plurality of rotations of said channel impulse response including and surrounding said a rotation that starts with coefficient L+1, where L is a length of said cyclic prefix; (2.2) calculating a value for inter-symbol interference based on each of said rotations; and (2.3) selecting a one of said rotations of said channel impulse response that yields the lowest inter-symbol interference value.
 13. The method of claim 11 wherein step (2.2) comprises calculating FINTF={overscore (C)}(|h′ _(Y)|(FV ₁ ·W)+|h′ _(Y+1)|(FV ₂ ·W)+ . . . +|h′ _(P−1)|(FV _(P−Y) ·W)) where {overscore (C)}=an average value of a transmitted discrete multitone symbol, Y=an integer selected based on the number of the next to last coefficient of the set of consecutive coefficients determined in step (3.1), P=the number of coefficients in said shortening channel impulse response, and W=is a weighting factor vector [w₀, w₁, w₂, . . . , w_(P−1)]^(T).
 14. A discrete multitone transceiver comprising: a transmitter; a receiver; a digital processing device adapted to generating a shortening channel impulse response by; determining an impulse response of a channel, said impulse response having a plurality of coefficients corresponding to a length of a symbol; and rotating said impulse response coefficients to a rotation that decreases inter-symbol interference value; and a timing recovery circuit that aligns with a received frame using said rotation.
 15. The transceiver of claim 14 wherein said digital processing device is adapted to determine said rotation by rotating said impulse response coefficients to a rotation in which the first L+1 coefficients of said channel impulse response is maximal.
 16. The transceiver of claim 14 wherein said digital processing device is adapted to determine said rotation by rotating said impulse response coefficients to a rotation that starts with coefficient L+1, where L is a length of said cyclic prefix.
 17. The transceiver of claim 14 wherein said digital processing device is adapted to determine said rotation by: selecting a plurality of rotations of said channel impulse response including and surrounding said a rotation that starts with coefficient L+1, where L is a length of said cyclic prefix; calculating a value for inter-symbol interference based on each of said rotations; and selecting a one of said rotations of said channel impulse response that yields the lowest inter-symbol interference value.
 18. The transceiver of claim 17 wherein said digital processing device performs said calculation by: generating Fourier transforms of said coefficients of said shortening channel impulse response; calculating an average value of a transmitted discrete multitone symbol; and multiplying said Fourier transforms of said coefficients with said average.
 19. The transceiver of claim 17 wherein said processor calculates said inter-symbol interference, FINTF, by; FINTF={overscore (C)}(|h′ _(Y)|(FV ₁ ·W)+|h′ _(Y+1)|(FV ₂ ·W)+ . . . +|h′ _(P−1)|(FV _(P−Y) ·W)) where C=an average value of a transmitted discrete multitone symbol, Y=an integer selected based on the number of the next to last coefficient of the set of consecutive coefficients determined in step (3.1), P=the number of coefficients in said shortening channel impulse response, and W=is a weighting factor vector [w₀, w₁, w₂, . . . , w_(P−1)]^(T).
 20. A method of frame alignment in a discrete multitone transceiver, said method comprising the steps of: (1) determining an impulse response of a channel, said impulse response having a plurality of coefficients corresponding to a length of a symbol; (2) determining a set of consecutive samples of said channel impulse response of length L+1, where L is a length of said cyclic prefix, for which the channel impulse response energy is maximal; (3) selecting a plurality of rotations of said shortening channel impulse response including and surrounding a rotation that starts with a first coefficient of said consecutive samples determined in step (3); (4) calculating a value for inter-symbol interference based on each of said rotations; and (5) selecting a one of said rotations selected is step (3) that decreases inter-symbol interference value.
 21. The method of claim 20 wherein step (4) comprises: (4.1) generating fast Fourier transforms of said coefficients of said channel impulse response; (4.2) calculating an average value of a transmitted discrete multitone symbol; and (4.3) multiplying said Fourier transforms of said coefficients with said average.
 22. The method of claim 20 wherein step (5) comprises calculating FINTF={overscore (C)}(|h′ _(Y)|(FV ₁ ·W)+|h′ _(Y+1)|(FV ₂ ·W)+ . . . +|h′ _(P−1)|(FV _(P−Y) ·W)) where C=an average value of a transmitted discrete multitone symbol, Y=an integer selected based on the number of the next to last coefficient of the set of consecutive coefficients determined in step (3.1), P=the number of coefficients in said shortening channel impulse response, and W=is a weighting factor vector [w₀, w₁, W₂, . . . , w_(P−1)]^(T). 